THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA

Tel: +1 508 478 9200; Fax: +1 508 478 0990; Web: www.thatcorp.com

THAT 4311

Description

The THAT 4311 Low Power Dynamics Proces-

sor combines in a single IC all the active circuitry
needed to construct a wide range of dynamics
processors.

The 4311 includes a high perfor-

mance, voltage controlled amplifier, a log re-
sponding RMS-level sensor and three opamps,
one of which is dedicated to the VCA, while the
other two may be used for the signal path or con-
trol voltage processing.

The exponentially-controlled VCA provides

two opposing-polarity, voltage sensitive control
ports. Dynamic range exceeds 105 dB, and THD
is typically 0.09% at 0dB gain. The RMS detector
provides accurate RMS to DC conversion over an

80 dB dynamic range.

Though originally designed for use in micro-

phone noise reduction systems, the 4311 is a use-
ful building block in a number of analog signal
processing applications. The combination of ex-
ponential VCA gain control and logarithmic detec-
tor

response

"decibel-linear"

response

simplifies the mathematics of designing the con-
trol paths of dynamics processors, making it easy
to develop audio compressors, limiters, gates, ex-
panders, de-essers, duckers, and the like.

The

high level of integration ensures excellent temper-
ature tracking between the VCA and the detector,
while minimizing the external parts count.

T H A T

C o r p o r a t i o n

Low-voltage, Low-power

Analog Engine

Dynamics Processor

FEATURES

High Performance VCA, RMS-Level
Detector, and three 0pamps in one
package

Wide Dynamic Range: >105 dB

Low THD: <0.09%

Low Power: 7 mA typ.

Surface-Mount Package

5 VDC Operation

APPLICATIONS

Wireless microphone systems

Wireless in-ear monitors

Compressors and Limiters

Gates

De-Essers

Duckers

THAT4311

VREF

OA1

OA2

VEE

EC-

EC+

SYM

OUT

VCA

RMS

OA3

VCC

VREF

Figure 1. THAT 4311 equivalent block diagram

Pin Name

DMP20

RMS IN

IT (I

TIME

)

OA2 -IN

RMS OUT

CT (C

TIME

)

CLIP

OA2 OUT

CAP

VREF

VEE

VCC

OA3 OUT

VCA OUT

SYM

EC+

EC-

VCA IN

OA1 OUT

OA1 -IN

OA1 +IN

Table 1. THAT 4311 pin assignments

THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA

Tel: +1 508 478 9200; Fax: +1 508 478 0990; Web: www.thatcorp.com

Page 2

Low-voltage, Low-power Analog Engine

Dynamics Processor

Absolute Maximum Ratings (T

= 25°C)

Positive Supply Voltage (V

)

+15 V

Power Dissipation (P

) (T

= 75°C)

700 mW

Operating Temperature Range (T

)

-20 to +70°C

Storage Temperature Range (T

)

-40 to +125°C

Max

C+

- (E

C-

)

± 1V

Electrical Characteristics

Parameter

Symbol

Conditions

Min

Typ

Max

Units

Supply Current

No signal; V

=+7 VDC

7.0

9.0

Reference Voltage

REF

1.8

1.95

2.1

Encode and Decode Companding Noise Reduction ( V

= +7V encoder, +15V decoder)

Encode Level Match

LMe

Encode mode; f = 1kHz

-25.3

-23.0

-20.7

dBV

Encode Gain Accuracy

Encode mode, f = 1kHz

GAe1

Vin = LMe + 10dB

+3.5

+6.5

GAe2

Vin = LMe - 40dB

-23

-20

-17

Decode Level Match

LMd

Decode mode; f = 1kHz

-18.3

-16.0

-13.7

Decode Gain Accuracy

Decode mode; f=1kHz

GAd1

Vin = LMd + 5dB

+8.5

+10

+11.5

GAd2

Vin = LMd - 20dB

-43

-40

-37

Max Input Voltage

Vime

Encode mode; THD = 3%; f = 1kHz

dBV

Max Output Voltage

Vomd

Decode mode; THD = 3%; f = 1kHz

10.7

13.7

dBV

Total Harmonic Distortion

THDtrim

End-to-end; Vin = LMe; f = 1kHz

0.025

(with trim)

Total Harmonic Distortion

THDnotrim

End-to-end; Vin = LMd; f = 1kHz

0.15

0.7

(no trim)

Output Noise

Vnod

End-to-end ; Vin = short; A-weighted

µVrms

Recommended Operating Conditions

Parameter

Symbol

Conditions

Min

Typ

Max

Units

Positive Supply Voltage

+15

SPECIFICATIONS

1. All specifications are subject to change without notice.
2. Unless otherwise noted, T

=25°C, test circuit as shown in Fig 2.

THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA

Tel: +1 508 478 9200; Fax: +1 508 478 0990; Web: www.thatcorp.com

Rev. 03/03/04

Page 3

Electrical Characteristics (con't)

Parameter

Symbol

Conditions

Min

Typ

Max

Units

Op amp OA1

Offset Voltage

RL = 2k

±0.5

±6

Equivalent Input Noise

nOA1

A-weighted

6.5

Total Harmonic Distortion

THD

OA1

1kHz, A

=1; R

= 10k

0.0007

0.003

Open Loop Gain

VO-OA1

= 10k

115

Gain Bandwidth Product

GBW

OA1

at 50kHz

Slew Rate

OA1

Op amp OA2

Offset Voltage

RL = 2k

±0.5

±6

Equivalent Input Noise

nOA1

A-weighted

7.5

Total Harmonic Distortion

THD

OA1

1kHz, A

=1; R

= 10k

0.0007

0.003

Open Loop Gain

VO-OA1

= 10k

110

Gain Bandwidth Product

GBW

OA1

at 50kHz

Slew Rate

OA1

R11

23k2

R14

31k6

R22

30R1

R16

280R

47u

V-

R12

261k

51R

300k 5%

20k0

47p NPO

20k0

47u

50k

Sym

V+

R10

100R 5%

10u

R26

100R

TP2

RMS Output

TP1

RMS Input

R17

31k6

C8 47p

R20

10k0

R21

10k0

3u3

CN1

External

Control

Input

CONTROL-VOLTAGE

CN2

Power

Input

C11
100n

(U1)

C10

22u

V+

Bypass

Capacitors

XLR2

XLR-M

Output

C17

47p

1N4004

OUT

VCA

EC+

SYM

EC-

OA3

VREF

U1A

THAT4311

Iset

OUT 4

RMS

U1B

THAT4311

OA2

VREF

U1C

THAT4311

20 OA1

U1D
THAT4311

CAP

VEE

VREF

VCC

VREF

U1E

THAT4311

OP-27

100k

1k33

10k0

R29

1k33 VREF

R23

280R

R24

2k80

C15

1000u

C16

47u

R27

10k0

R18

10k0

R19

100k

V+

R15

31k6

C19
1000u

10u

22u

10u

V+

VREF

10k0

C12
3u3

+40dB

+20dB

+40dB

+60dB

C18
47u

SW1B

SW1E

SW1F

C14

100n

(U1)

C13

22u

1N4004

V-

VREF

R25

100R

R28

100R 5%

SW1C

SW1A

CN1

RMS Output

VREF

0dB

5 4

3 1

XLR1

XLR-F

Input

Fig 2. THAT 4311 test circuit

THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA

Tel: +1 508 478 9200; Fax: +1 508 478 0990; Web: www.thatcorp.com

Page 6

Low-voltage, Low-power Analog Engine

Dynamics Processor

Theory of Operation

The THAT 4311 Analog Engineâ Dynamics Pro-

cessor combines THAT,s proven Voltage-Controlled
Amplifier (VCA) and RMS-Level Detector designs with
three opamps to produce a multipurpose dynamics
processor useful in a variety of applications. For de-
tails of the theory of operation of the VCA and RMS
Detector building blocks, the interested reader is re-
ferred to THAT Corporation's data sheets on the
218x Series VCAs and the 2252 RMS-Level Detector.
Theory

the

interconnection

exponen-

tially-controlled VCAs and log-responding level detec-
tors is covered in THAT Corporation's application
note AN101, The Mathematics of Log-Based Dynamic
Processors.

The VCA - in Brief

The THAT 4311 VCA is based on THAT Corpora-

tion's highly successful complementary log/anti-log
gain cell topology, as used in THAT's 218x and
215x-Series IC VCAs. The THAT 4311 is integrated
using a fully complementary, BiFET process.

The

combination of FETs with high-quality, complemen-
tary bipolar transistors (NPNs and PNPS) allows ad-
ditional flexibility in the design of the VCA over
previous efforts.

Input signals are currents to the VCA IN pin.

This pin is a virtual ground biased at VREF, so in
normal operation an input voltage is converted to in-
put current via an appropriately sized resistor (R5 in
Fig 2, Page 3). Because the current associated with
DC offsets relative to VREF present at the input pin
and any DC offset in preceding stages will be modu-
lated by gain changes (thereby becoming audible as
thumps), the input pin is normally AC-coupled (C4 in
Fig 2).

The VCA output signal is also a current, inverted

with respect to the input current. In normal opera-
tion, the output current is converted to a voltage via
inverter OA3, where the ratio of the conversion is de-
termined by the feedback resistor (R6, Fig 2) con-
nected between OA3's output and its inverting input.
The signal path through the VCA and OA3 is
non-inverting.

The gain of the VCA is controlled by the voltage

applied between EC- and the combination of EC+
and SYM. Gain (in decibels) is proportional to EC-,
provided that EC+ and SYM are at VREF. The con-
stant of proportionality is -6.1mV/dB (for 5V sup-
plies) for the voltage at EC-, and 6.1mV/dB for the
voltage at EC+, and SYM

OUT

VCA

EC+

SYM

EC-

OA3

Vref

U1A

THAT4311

Iset

OUT

RMS

U1B

THAT4311

Cap

Vee

Vref

Vcc

Vref

U1E

THAT4311

24k3

R9
51R

150k

24k3

R11

15k

R12

261k

3n3

3u3

10u

20k

10u

V+

C2
10u

C3
22u

Decoder In

Decoder Out

6k04

+15

56k

8k87

C16

10u

OA1

U1D
THAT4311

7k5

10u

Fig 16. THAT 4311 Noise Reduction Decoder Schematic

THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA

Tel: +1 508 478 9200; Fax: +1 508 478 0990; Web: www.thatcorp.com

Rev. 03/03/04

Page 7

As mentioned, for proper operation, the same

voltage must be applied to EC+ and SYM, except for
a small (±2.5 mV) DC bias applied between these
pins.

This bias voltage adjusts for internal mis-

matches in the VCA gain cell which would otherwise
cause small differences between the gain of positive
and negative half-cycles of the signal. The voltage is
usually applied via an external trim potentiometer
(R7 in Fig 2), which is adjusted for minimum signal
distortion at unity (zero dB) gain.

The VCA may be controlled via EC-, as shown in

Fig 17, or via the combination of EC+ and SYM.
This connection is illustrated in Fig 18.

Note that

this latter figure shows only that portion of the cir-
cuitry needed to drive the positive VCA control port;
circuitry associated with OA1, OA2 and the RMS de-
tector has been omitted.

While the 4311's VCA circuitry is very similar to

that of the THAT 2180 Series VCAs, there are several
important differences, as follows:

1. Supply current for the VCA is fixed internally.

Approximately 500

mA is available for the sum of in-

put and output signal currents.

2. The signal current output of the VCA is inter-

nally connected to the inverting input of an on-chip

opamp.

In order to provide external feedback

around this opamp, this node is brought out to a pin.

3. The input stage of the 4311 VCA uses inte-

grated P-channel FETs rather than a bias-current
corrected bipolar differential amplifier.

Input bias

currents have therefore been reduced.

The RMS Detector - in Brief

The THAT 4311's detector computes RMS level

by rectifying input current signals, converting the rec-
tified current to a logarithmic voltage, and applying
that voltage to a log-domain filter. The output signal
is a DC voltage proportional to the decibel-level of the
RMS value of the input signal current.

Some AC

component (at twice the input frequency) remains su-
perimposed on the DC output. The AC signal is at-
tenuated by a log-domain filter, which constitutes a
single-pole roll-off with cutoff determined by an ex-
ternal capacitor and a programmable DC current.

As in the VCA, input signals are currents to the

RMS IN pin. This input is a virtual ground biased at
VREF, so a resistor (R11 in Fig 2) is normally used to
convert input voltages to the desired current.

The

level detector is capable of accurately resolving sig-

nals well below 10mV (with a 10k

W input resistor).

However, if the detector is to accurately track such
low-level signals, AC coupling is normally required.

OUT

VCA

EC+

SYM

EC-

OA3

Vref

U1A

THAT 4311

Iset

OUT

RMS

U1B

THAT 4311

OA2

Vref

U1C

THAT 4311

OA1

U1D

THAT 4311

Cap

Vee

Vref

Vcc

Vref

U1E

THAT 4311

Vref

20k

51R

51k

10k

264k

47u

47p

47u

10u

22u

100n

20k

Signal In

RMS Out

Signal Out

10u

Control Port Drive

Fig 17. Circuit showing gain control at E

C-

THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA

Tel: +1 508 478 9200; Fax: +1 508 478 0990; Web: www.thatcorp.com

Page 8

Low-voltage, Low-power Analog Engine

Dynamics Processor

The log-domain filter cutoff frequency is usually

placed well below the frequency range of interest.
For an audio-band detector, a typical value would be

5Hz, or a 32ms time constant (

t). The filter's time

constant is determined by an external capacitor at-
tached to the CT pin, and an internal current source
(I

TIME

) connected to CT. The current source is pro-

grammed via the IT pin: current in IT is mirrored to
I

TIME

with a gain of approximately one. The resulting

time

constant

t is approximately equal to

(0.026

´ CT) / I

. Note that, as a result of the mathe-

matics of rms detection, the attack and release time
constants are fixed in their relationship to each other.

The DC output of the detector is scaled with the

same constant of proportionality as the VCA gain
control: 6.1mV/dB. The detector's zero dB reference
(Iin0, the input current which causes zero volts out-
put), is determined by IT as follows:

Iin0=IT. The

detector output stage is capable of sinking or sour-

cing l00

mA.

Differences between the 4311's RMS-Level Detec-

tor circuitry and that of the THAT 2252 RMS Detec-
tor are as follows:

1. The rectifier in the 4311 RMS Detector is inter-

nally balanced by design, and cannot be balanced via
an external control. The 4311 will typically balance
positive and negative halves of the input signal within

±1.5%, but in extreme cases the mismatch may
reach +20%. However, a 20% mismatch will not sig-
nificantly increase ripple-induced distortion in dy-
namics processors over that caused by signal ripple
alone.

2. The time constant of the 4311's RMS detector

is determined by the combination of an external ca-
pacitor (connected to the CT pin) and an internal,
programmable current source. The current source is
equal to IT. Normally, a resistor is not connected di-
rectly to the CT pin on the 4311.

3. The zero dB reference point, or level match, is

not adjustable via an external current source. How-
ever, as in the 2252, the level match is affected by the
timing current, which, in this case, is drawn from the
IT pin and mirrored internally to CT.

4. The input stage of the 4311 RMS detector uses

integrated P-channel FETs rather than a bias-current
corrected bipolar differential amplifier.

Input bias

currents are therefore negligible, improving perfor-
mance at low signal levels.

The Opamps - in Brief

The three opamps in the 4311 are intended for

general purpose applications. All are 5MHz opamps

with slew rates of approximately 2V/

ms. All use bipo-

lar PNP input stages. However, the design of each is
optimized for its expected use. Therefore, to get the

OUT

VCA

EC+

SYM

EC-

OA3

Vref

U1A

THAT 4311

Iset

OUT

RMS

U1B

THAT 4311

OA2

Vref

U1C

THAT 4311

OA1

U1D

THAT 4311

Cap

Vee

Vref 9

Vcc

Vref

U1E

THAT 4311

Vref

20k

51R

51k

47u

47p

10u

22u

100n

50k

Signal In

Signal Out

Control Port Drive

N/C

Fig 18. Circuit showing gain control at E

C+

THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA

Tel: +1 508 478 9200; Fax: +1 508 478 0990; Web: www.thatcorp.com

Rev. 03/03/04

Page 9

most out of the 4311, it is useful to know the major
differences among these opamps.

OA3, being internally connected to the output of

the VCA. is intended for current-to-voltage conver-
sion.

Its input noise performance, at 7 5

Hz,

complements that of the VCA, adding negligible noise
at unity gain. Its output section is capable of driving

1mA into a 2k

W load.

OA1 is the quietest opamp of the three, and with

its typical input referred noise of 6 5

Hz, is the

opamp of choice for input stages. Its output section

is nominally capable of driving 3mA into a 5k

W load.

OA2 is best suited for control voltage processing,

though is does have anti-paralleled diodes that can

be used to fashion it into a clipper. (However in most
applications where a clipper is needed, it's preferable
to place it around OA3). OA2's input noise is compa-
rable to OA3 and its output drive is comparably to
OA1.

The Reference Voltage - In Brief

THAT Corporation's log/anti-log VCAs and RMS`

detectors require a reference voltage between the pos-
itive and negative power supplies, and to supply this,
the THAT 4311 provides an on-chip, 2V reference
about which the VCA, the RMS detector, and OA2 are
biased. This reference is a buffered band-gap refer-
ence that is amplified to 2V. Pins are provided for fil-
ter capacitors at both the input and the output of the
buffer, which are labeled CAP and VREF respectively.

Application Information

As noted previously, the THAT 4311 was origi-

nally designed for noise reduction systems, hence the
emphasis on those parameters in the specifications.
Its low power consumption, integration, and similar-
ity to the THAT 4301, however, extend its utility to a
variety of other products and applications. The cir-
cuit of Fig 19, shows a typical application for the
THAT 4311. This simple compressor/limiter design
features adjustable hard-knee threshold, compres-

sion ratio, and static gain. The applications discus-
sion in this data sheet will center on this circuit for
the purpose of illustrating important design issues.

Signal Path

As mentioned in the section on theory, the VCA

input pin is a virtual ground with negative feedback

provided internally. An input resistor (R1, 20k

W) is

required to convert the AC input voltage to a current

OUT

VCA

EC+

SYM

EC-

OA3

Vref

U1A

THAT4311

Iset

OUT

RMS

U1B

THAT 4311

OA2

Vref

U1C

THAT4311

OA1

U1D

THAT4311

Cap

Vee

Vref

Vcc

Vref

U1E

THAT 4311

20k

51R

33k

20k

R15

10k

28k7

267k

4k99

10k

R10

82k

R12
20k

R13
10k

R11

51k

R17

82k

R16

4k99

R14

1k43

R18

50k

47u

47p

47u

10u

100n

10u

C9
22u

Vref

10u

Out

Vin0=-10dBu

CR1

1N4148

CR2

1N4148

Gain

Threshold

Compression

Ratio

Fig 19. Simple compressor / limiter using the THAT 4311

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Tel: +1 508 478 9200; Fax: +1 508 478 0990; Web: www.thatcorp.com

Page 10

Low-voltage, Low-power Analog Engine

Dynamics Processor

within the linear range of the THAT 4311. (Peak VCA

input currents should be kept under 250

mA for best

distortion performance.) The coupling capacitor (Cl,

mF) is strongly recommended to block DC current

from preceding stages (and from offset voltage at the
input of the VCA). Any DC current into the VCA will
be modulated by varying gain in the VCA, showing up
in the output as "thumps". Note that Cl, in conjunc-
tion with R1, will set the low frequency limit of the
circuit.

The VCA output is connected to OA3, configured

as an inverting current-to-voltage converter.

OA3's

feedback components (R2, 20 k

W, and C2, 47 pF) de-

termine the constant of current-to-voltage conversion.
The simplest way to deal with this is to recognize that
when the VCA is set for unity (zero dB) gain, the in-
put to output voltage gain is simply R2/R1, much like
the case of a single inverting stage. If, for some rea-
son, more than zero dB gain is required when the
VCA is set to unity, then the resistors may be skewed
to provide it. Note that the feedback capacitor (C2) is
required for stability. The VCA output has approxi-
mately 45pF of capacitance to ground, which must be
neutralized via the 47pF feedback capacitor across
R2.

The VCA gain is controlled via the EC- terminal,

whereby gain in dB will be proportional to the nega-
tive of the voltage at EC-. In this application the EC+
terminal is tied to VREF, though it could be the
driven port, or the control ports could be driven dif-
ferentially. The SYM terminal is returned nearly to
the EC+ terminal (which is in this case VREF) via a

small resistor (R3, 51

W). The VCA SYM trim (R5,

20k

W) allows a small voltage to be applied to the

SYM terminal via R4 (33k

W). This voltage adjusts for

small mismatches within the VCA gain cell, thereby
reducing even-order distortion products.

To adjust

the trim, apply to the input a middle-level, mid-
dle-frequency signal (1kHz at 200mV

rms

is a good

choice with this circuit) and observe THD at the sig-
nal output. Adjust the trim for minimum THD.

RMS-Level Detector

The RMS detector's input is similar to that of the

VCA. An input resistor (R6, 28.7k

W) converts the AC

input voltage to a current within the linear range of

the THAT 4311. The coupling capacitor (C3, 47

mF)

is recommended to block the current from preceding
stages (and from offset voltage at the input of the de-
tector). Any DC current into the detector will limit
the low-level resolution of the detector, and will upset
the rectifier balance at low levels. Note that, as with

the VCA input circuitry, C3 in conjunction with R6
will set the lower frequency limit of the detector.

The time response of the RMS detector is deter-

mined by the capacitor attached to CT (C4, 10 uF)
and the size of the current in pin IT (determined by

R7, 267 k

W and VREF, 2V). Since the voltage at IT is

approximately 2V, the circuit of Figure 19 produces

7.5

mA in IT, The current in IT is mirrored to the CT

pin, where it is available to discharge the timing ca-
pacitor (C4). The combination produces a log filter
with time constant equal to approximately 0.026
CT/IT (~35 ms in the circuit shown).

The waveform at CT will follow the logged (deci-

bel) value of the input signal envelope, plus a DC off-
set of about 2V

plus VREF or about 3.3V.

The

capacitor used should be a low-leakage, electrolytic
type in order not to add significantly to the timing
current.

The output stage of the RMS detector serves to

buffer the voltage at CT and removes the 1.3 V

(2 V

) offset, resulting in an output centered around

VREF for input signals of about 245 mV

rms

, or

-10 dBu. The output voltage increases 6.1 mV for ev-
ery 1 dB increase in input signal level. This relation-
ship holds over more than a 60 dB range in input
currents.

Control Path

The primary function of an audio compressor is

to reduce a signal's dynamic range. A 2:1 compres-
sor reduces a 100 dB dynamic range to 50 dB.

limiter, or infinite compressor, is a special case of
compressor where the dynamic range is reduced to
the point where the rms level of the signal is con-
stant.

This reduction in dynamic range is accom-

plished by a) raising the gain when the signal is
below some particular level -- often referred to as the
'zero dB reference level' -- and b) reducing the gain
when the signal is above that level. In addition, these
devices often have a threshold, below which the sig-
nal is passed unprocessed and above which compres-
sion takes place. This feature keeps the noise floor
from rising to noticeable levels in the absence of sig-
nal.

We previously established that the zero dB refer-

ence level of the detector is -10 dBu (zero dB refer-

ence level = 7.5

mA , R6 = 28.7 kilohms). Neglecting

the effect of the threshold control (R11 and R12),
when the output is below this level the output of OA2
is driven high, forward biasing CR1 and reverse bias-
ing CR2. Since CR2 is not conducting, no signal is
passed to the VCA's control port by OA1. When the
signal level exceeds -10 dBu, the output of the RMS
detector goes positive, and CR2 begins to conduct. In

THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA

Tel: +1 508 478 9200; Fax: +1 508 478 0990; Web: www.thatcorp.com

Rev. 03/03/04

Page 11

this case, OA2's feedback is provided through R9,
and the sensitivity at this point is 12.2 mV/dB, since
the output of the RMS detector is multiplied by
-R9/R8, or a gain of -2.

A threshold control is provided to vary the

threshold above or below -10 dBu. The output sensi-
tivity of the RMS detector is 6.1 mV/dB. This is con-
verted to a current by R8, and the sensitivity at the
summing node of OA2 is

6 1

4 99

The wiper of R12 can swing between -2V and +3V

relative to the summing node of OA2 which is at
VREF. If we want the threshold to swing as high as
+30 dB, then the value required for R11 can be cal-
culated as

1 2

when rounded to the nearest 5% resistor value.

Using this value and knowing that the pot's swing in
the other direction is 3V, we can calculate the thresh-
old swing in the negative direction to be

1 2

V
k

» -

Since the zero dB reference level of the detector is

-10 dBu, the threshold can be adjusted from 20 dBu
to -59 dBu.

The output of the threshold detector represents

the signal level above the determined threshold, at a
constant

about

13mV/dB

(from

[R9/R8]

6.1mV/dB).

This signal is passed on to the COM-

PRESSION control (R13), which variably attenuates
the signal passed on to OA1. Note that the gain of
OA1, from the wiper of the COMPRESSION control to
OA1's output is R16/R15 (0.5), precisely the inverse
of the gain of OA2.

Therefore, the COMPRESSION

control lets the user vary the above threshold gain be-
tween the RMS detector output and the output of
OA1, from zero to a maximum of unity.

The gain control constant of the VCA (6.1mV/dB)

is exactly equal to the output scaling constant of the
RMS detector.

Therefore, at maximum COMPRES-

SION, above threshold, every dB increase in input
signal level causes a 6.1mV increase in the output of
OA1, which in turn causes a 1dB decrease in the VCA
gain. With this setting, the output will not increase
despite large increases in input level above threshold.
This is infinite compression.

For intermediate set-

tings of COMPRESSION, a 1dB increase in input sig-
nal level will cause less than a 1dB decrease in gain,
thereby varying the compression ratio.

The resistor R14 is included to alter the taper of

the COMPRESSION pot to better suit common usage.

If a linear taper pot is used for R13, the compression
ratio will be 1:2 at the middle of the rotation. How-
ever, 1:2 compression in an above-threshold com-
pressor is not very strong processing, so 1:4 is often
preferred at the midpoint.

R14 warps the taper of

R13 so that 1:4 compression occurs at approximately
the midpoint of R13's rotation,

The GAIN control (R18) is used to provide static

gain or attenuation in the signal path. This control
adds between 120mV and -180mV of offset to the
output of OA1, which is approximately a -20dB to
+30dB change in gain of the VCA. The gain control
signal is passed into OA1 via R17, but this signal is
also passed back to the threshold amplifier (OA2) via
R10. This arrangement results in the threshold be-
ing fixed relative to the output. In other words, as
the gain is increased, the threshold is lowered to
keep the threshold of compression or limiting at the
same output level. This is particularly important in
limiters, since it keeps the gain control from interact-
ing with the threshold.

C5 is used to attenuate the noise of OA1, OA2,

and the resistors R8 through R16 used in the control
path. All these active and passive components pro-
duce noise which is passed on to the control port of
the VCA, causing modulation of the signal. By itself,
the THAT 4311 VCA produces very little noise modu-
lation, and its performance can be significantly de-
graded by the use of noisy components in the control
voltage path.

Overall Result

The

resulting

compressor

circuit

provides

hard-knee compression above threshold with three
essential user adjustable controls. The threshold of
compression may be varied over a range from about
-58dBu to +20dBu. The compression ratio may be

varied from 1:1 (no compression) to

¥:1. And, static

gain may be added between -20 and +30dB. Audio
performance is excellent, with THD running below
0.1% at middle frequencies even with 10 dB of com-
pression, and an input dynamic range of over 105dB.

Perhaps most important, this example design

only scratches the surface of the large body of appli-
cations circuits which may be constructed with the
THAT 4311. The combination of an accurate, wide
dynamic range, log-responding level detector with a
high-quality, exponentially-responding VCA produces
a versatile and powerful analog engine. These, along
with its on-board opamps, allow a designer to con-
struct, with a single IC and a handful of external
components, gates, expanders, de-essers, noise re-
duction systems and the like.

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